The present invention relates to communication methods and electronic devices, and, more particularly, to communication methods and electronic devices that use automatic frequency control (AFC) systems and methods.
Wireless communications systems are commonly used to provide voice and data communications to subscribers. For example, analog cellular radiotelephone systems, such as those designated AMPS, ETACS, NMT-450, and NMT-900, have long been deployed successfully throughout the world. Digital cellular radiotelephone systems such as those conforming to the North American standard IS-54 and the European standard GSM have been in service since the early 1990's. More recently, a wide variety of wireless digital services broadly labeled as PCS (Personal Communications Services) have been introduced, including advanced digital cellular systems conforming to standards such as IS-136, IS-95, and UMTS, lower-power systems such as DECT (Digital Enhanced Cordless Telephone) and data communications services such as CDPD (Cellular Digital Packet Data). These and other systems are described in The Mobile Communications Handbook, edited by Gibson and published by CRC Press (1996).
Several types of access techniques are conventionally used to provide wireless services to users of wireless systems. Traditional analog cellular systems generally use a system referred to as Frequency Division Multiple Access (FDMA) to create communications channels, wherein discrete frequency bands serve as channels over which cellular terminals communicate with cellular base stations. Typically, these bands are reused in geographically separated cells in order to increase system capacity.
Modern digital wireless systems typically use different multiple access techniques such as Time Division Multiple Access (TDMA) and/or Code Division Multiple Access (CDMA) to provide increased spectral efficiency. In TDMA systems, such as those conforming to the GSM or IS-136 standards, carriers are divided into sequential time slots that are assigned to multiple channels such that a plurality of channels may be multiplexed on a single carrier. CDMA systems, such as those conforming to the IS-95 and UMTS standards, achieve increased channel capacity by using “spread spectrum” techniques wherein a channel is defined by modulating a data-modulated carrier signal by a unique spreading code, i.e., a code that spreads an original data-modulated carrier over a wide portion of the frequency spectrum in which the communications system operates. The spreading code typically includes a sequence of “chips” occurring at a chip rate that is higher than the bit rate of the data being transmitted.
A so-called RAKE receiver structure is commonly used to recover information corresponding to one of the user data streams. In a typical RAKE receiver, a received composite signal is correlated with a particular spreading sequence assigned to the receiver to produce a plurality of time-offset correlations, a respective one of which corresponds to an echo of a transmitted spread spectrum signal. The correlations are then combined in a weighted fashion, i.e., respective correlations are multiplied by respective weighting factors and then summed to produce a decision statistic. The correlations are generally performed in a plurality of correlating fingers in the RAKE receiver, wherein each finger is synchronized with a channel path. The outputs of all fingers are combined to allow an improvement in the overall signal-to-noise ratio of the received signal. The design and operation of RAKE receivers are well known to those having skill in the art and need not be described further herein.
To maintain the RAKE receiver fingers synchronized with their respective channel paths, a path searcher may be used to support the RAKE receiver. The path searcher can continuously search for new channel paths and estimate their delays. These delays are then assigned to the RAKE fingers. For a wideband CDMA (WCDMA) system, the detection of the multi-path delays is typically done as a two-stage process: In the first stage, a wide search is done to identify the location of the multi-path delays. The resolution of this first search (i.e., the separation between the delays) is typically one chip or less. Typically, the received power or signal to interference ratio (SIR) is used as a criterion for the quality of the delayed signal. In the second stage, a localized search is performed over selected regions of delays. The resolution of this second search is typically one-half chip to an eighth of a chip. A decision is then made as to which delays to use for despreading the data based on the information from the localized search.
Referring now to FIG. 1, a conventional terrestrial cellular radiotelephone communication system 20 is illustrated. The cellular radiotelephone communication system 20 may include one or more radiotelephones (terminals) 22, communicating with a plurality of base stations 26 serving a plurality of cells 24 and a mobile telephone switching office (MTSO) 28. Although only three cells 24 are shown in FIG. 1, a typical cellular network may include hundreds of cells, may include more than one MTSO, and may serve thousands of radiotelephones.
The cells 24 generally serve as nodes in the communication system 20, from which links are established between radiotelephones 22 and the MTSO 28, by way of the base stations 26 serving the cells 24. Each cell 24 will have allocated to it one or more dedicated control channels and one or more traffic channels. A control channel is a dedicated channel used for transmitting cell identification and paging information. The traffic channels carry the voice and data information. Through the cellular network 20, a duplex radio communication link may be established between two mobile terminals 22 or between a mobile terminal 22 and a landline telephone user 32 through a Public Switched Telephone Network (PSTN) 34. The function of a base station 26 is to handle radio communication with mobile terminals 22 within the cells 24. In this capacity, a base station 26 functions as a relay station for data and voice signals.
It is generally desirable to ensure good frequency synchronization between a mobile terminal and a base station. Thus, a mobile terminal may include an automatic frequency control (AFC) block or component to keep the frequency difference between a base station and a mobile terminal within acceptable requirements for the system in question. In a Universal Mobile Telephone System (UMTS) using WCDMA, the frequency accuracy of the mobile terminal transmitter is specified to be within 0.1 ppm of a received base-station frequency, for example, around 200 Hz for a system operating at 2 GHz.
Referring now to FIG. 2, a conventional mobile terminal architecture 200 that includes an AFC component comprises a transceiver 210, an AFC-algorithm component 220, a conversion component 230, a digital-to-analog converter (DAC) 240, and a voltage controlled crystal oscillator (VCXO) 250, which are configured as shown. The AFC-algorithm component 220 generates an error signal responsive to an incoming signal received through the transceiver 210. The error signal is converted by the conversion block 230 into an appropriate digital adjustment signal for the DAC 240, which generates a new output voltage to adjust the frequency generated by the VCXO 250.
A mobile terminal may use different AFC algorithms depending on whether the frequency error is expected to be relatively large or small. Referring now to FIG. 3, a conventional AFC-algorithm component 300 that may be used when the frequency error is expected to be relatively small comprises averaging blocks 310a, 310b, and 310c respectively associated with fingers of a RAKE receiver, delay operators 320a, 320b, and 320c, multipliers 330a, 330b, and 330c, summation component 340, an infinite impulse response (IIR) low pass filter 350, a conversion component 360, a multiplier 370, and an update decision component 380, which are configured as shown. In FIG. 3, S denotes received symbols, Sref denotes known pilot symbol references, and N is the number of symbols processed. As shown in FIG. 3, frequency error estimates are calculated for each RAKE receiver finger and combined by the summation component 340. The low pass filter 350 may be used to decrease the influence of Doppler variations in the frequency error. The conversion component 360 generates a frequency error signal by computing the arctan of the real portion of the signal output from the low pass filter 350 divided by the imaginary portion of the signal output from the low pass filter 350. The multiplier 370 is used to scale the frequency error signal output from the conversion component 360 with the signal fres to generate an error signal fe that may be used by the update decision component 380 to change the output voltage generated by a DAC for controlling a VCXO. In other implementations, separate frequency error measurements may be obtained from different base stations and a mean frequency error may be formed by weighting the frequency error measurements from the various base stations differently.
Referring now to FIG. 4, a conventional AFC-algorithm component 400 that may be used when the frequency error is expected to be relatively large comprises a read pilot symbol component 410, an estimation component 420, a zero-pad component 430, a fast fourier transform (FFT) component 440, an absolute value component 450, a summation component 460, a conversion component 470, a multiplier component 480, an update decision component 490, a frequency storage component 492, a summation component 494, and a comparator 496, which are configured as shown. For each slot, the read pilot symbol component 410 reads the pilot symbols which are used by the estimation component 420 to generate estimates where S denotes received symbols, Sref denotes known pilot symbol references, and * denotes complex conjugation. Components 430, 440, and 450 are used to generate a FFT of the estimates to convert to the frequency domain to generate an output that corresponds to the magnitude of the FFT squared. These values are summed by the summation component 460 and provided to the conversion component 470, which generates a frequency error signal by interpolating between adjacent frequencies associated with the maximum energy levels.
The multiplier 480 is used to scale the frequency error signal output from the conversion component 470 with the signal fres to generate an error signal fe that may be used by the update decision component 490 to change the output voltage generated by a DAC for controlling a VCXO. To determine whether to adjust the input signal to the DAC, the frequency storage component 492 stores the frequency that is associated with the maximum power level. The summation component 494 adds the power levels for the frequency associated with the maximum power and an adjacent and/or proximal frequency and provides this sum to the comparator 496. If the sum exceeds a threshold, then the update decision component 490 will adjust the input signal to the DAC so as to change the frequency generated by the VCXO.
Unfortunately, a DAC generally has limited precision. The resolution is typically equal to the analog-value range divided by 2n bits, where n is the number of bits used for the digital input signal. The frequency resolution may be approximately 50–100 Hz in some implementations. This large quantization may lead to sudden frequency changes when the input signal to the DAC changes. In addition, a VCXO may experience frequency drift in response to temperature changes. The temperature may change, for example, when the mode of a user terminal (i.e., the on-off status of the mobile terminal) changes and/or when the mobile terminal processor load changes.